Semiconductor device

ABSTRACT

A semiconductor device according to an aspect of the invention relates to an AD converter that converts a signal level of an analog signal into a digital value by using a comparator, and determines an amount of adjustment of an offset voltage of the comparator based on an offset determination result of the comparator obtained immediately after a least significant bit (LSB) of a digital value output as a conversion result is converted.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is based upon and claims the benefit of priority fromJapanese patent application No. 2013-258486, filed on Dec. 13, 2013, thedisclosure of which is incorporated herein in its entirety by reference.

BACKGROUND

The present invention relates to a semiconductor device, and moreparticularly, to a semiconductor device including, for example, ananalog-to-digital conversion circuit including a comparator.

Successive approximation AD (Analog-to-Digital) converters include twotypes, i.e., a synchronous operation type and an asynchronous operationtype. Synchronous operation type successive approximation AD converters,which perform a successive approximation operation in synchronizationwith an external clock signal, require a high-frequency external clocksignal to operate at a high speed. On the other hand, asynchronousoperation type successive approximation AD converters internallygenerate a clock signal based on an externally-input clock signal,thereby making it possible to operate at a higher speed than in the caseof using the external clock signal. For this reason, the asynchronousoperation type successive approximation AD converters are used forhigh-accuracy and high-speed AD conversion. The asynchronous operationtype successive approximation AD converters can be applied to, forexample, wireless communication devices and industrial equipment.

In a successive approximation AD converter called a charge-sharing typesuccessive approximation AD converter, the accuracy of the AD conversiondeteriorates due to an offset of a comparator. Accordingly, it isnecessary to perform not only the successive approximation operation,but also an offset correction operation to correct the comparator offsetto zero. However, it is difficult for applications requiring constantoperation, such as wireless communications (WCDMA®, FD-LTE, etc.) androtary encoders intended for industrial equipment, to perform thecorrection operation for correcting the comparator offset. It is alsopossible to employ a method for correcting the comparator offset onlyonce before use (for example, foreground operation). However, thecomparator offset varies when there is a change in the environmentduring use, which leads to deterioration in the accuracy of ADconversion.

In this regard, Japanese Unexamined Patent Application Publication No.2007-259224 discloses a technique for correcting an offset of acomparator to zero by a background operation. In an AD converterdisclosed in Japanese Unexamined Patent Application Publication No.2007-259224, a comparator input is short-circuited during sampling and acomparator output is held in a capacitor, to thereby correct the offsetof the comparator.

SUMMARY

However, the AD converter disclosed in Japanese Unexamined PatentApplication Publication No. 2007-259224 is a static operation type ADconverter that holds the output value of the comparator in thecapacitor. Therefore, the correction method cannot be applied to adynamic operation type AD converter that performs a conversion operationin synchronization with a clock signal. In other words, the related arthas a problem that the type of AD converters capable of correcting anoffset is limited.

The other problems to be solved and novel features of the invention willbecome apparent from the following description and the attacheddrawings.

A semiconductor device according to a first aspect of the presentinvention relates to an AD converter that converts a signal level of ananalog signal into a digital value by using a comparator, and determinesan amount of adjustment of an offset voltage of the comparator based onan offset determination result of the comparator obtained immediatelyafter a least significant bit (LSB) of a digital value output as aconversion result is converted.

According to the aspect of the invention, the semiconductor device canreduce the amount of offset of various types of AD converters.

BRIEF DESCRIPTION OF THE DRAWINGS

The above and other aspects, advantages and features will be moreapparent from the following description of certain embodiments taken inconjunction with the accompanying drawings, in which:

FIG. 1 is a block diagram showing an AD converter according to a firstembodiment;

FIG. 2 is a circuit diagram showing a sample-and-hold circuit accordingto the first embodiment;

FIG. 3 is a timing diagram for explaining non-overlapping clock signalsused for the AD converter according to the first embodiment;

FIG. 4 is a diagram for explaining an operation of the sample-and-holdcircuit according to the first embodiment;

FIG. 5 is a circuit diagram showing a DAC according to the firstembodiment;

FIG. 6 is a diagram for explaining an operation of the DAC according tothe first embodiment;

FIG. 7 is a circuit diagram showing a comparator according to the firstembodiment;

FIG. 8 is a timing diagram for explaining an operation of the comparatoraccording to the first embodiment;

FIG. 9 is a timing diagram for explaining an operation of the comparatoraccording to the first embodiment;

FIG. 10 is a timing diagram for explaining an operation of thecomparator according to the first embodiment;

FIG. 11 is a timing diagram for explaining an operation of the ADconverter according to the first embodiment;

FIG. 12 is a timing diagram for explaining values indicated by a DCOCcomparison determination result signal and an offset adjustment amountin the AD converter according to the first embodiment;

FIG. 13 is a schematic diagram for explaining a layout of the comparatoraccording to the first embodiment;

FIG. 14 is a schematic diagram for explaining a layout of thecomparator, a switch circuit, and lines according to the firstembodiment;

FIG. 15 is a schematic diagram for explaining in detail a layout of theswitch circuit and comparison lines according to the first embodiment;

FIG. 16 is a schematic diagram for explaining a positional relationshipbetween the switch circuit and the comparison lines in the layoutaccording to the first embodiment;

FIG. 17 is a timing diagram for explaining values indicated by a DCOCcomparison determination result signal and an offset adjustment amountin an AD converter according to a second embodiment;

FIG. 18 is a graph for explaining another example of the valuesindicated by the DCOC comparison determination result signal and theoffset adjustment amount in the AD converter according to the secondembodiment;

FIG. 19 is a graph for explaining a relationship between an updatetiming of an offset control value and the number of integrations of thevalues indicated by the DCOC comparison determination result signal inthe AD converter according to the second embodiment;

FIG. 20 is a graph for explaining another example of the relationshipbetween the update timing of the offset control value and the number ofintegrations of the values indicated by the DCOC comparisondetermination result signal in the AD converter according to the secondembodiment; and

FIG. 21 is a graph for explaining further another example of therelationship between the update timing of the offset control value andthe number of integrations of the values indicated by the DCOCcomparison determination result signal in the AD converter according tothe second embodiment.

DETAILED DESCRIPTION

The following description and the drawings are omitted and simplified asappropriate to clarify the explanation. The elements illustrated in thedrawings as functional blocks for performing various processes can beimplemented hardwarewise by a CPU, a memory, and other circuits, and canbe implemented softwarewise by a program loaded into a memory.Accordingly, it is understood by those skilled in the art that thefunctional blocks can be achieved in various forms including hardwarealone, software alone, and combinations thereof, and are not limited toany of them. Note that like elements are designated by identicalreference numerals throughout the drawings, and the description thereofis omitted as necessary.

The above-mentioned program can be stored and provided to a computerusing any type of non-transitory computer readable media. Non-transitorycomputer readable media include any type of tangible storage media.Examples of non-transitory computer readable media include magneticstorage media (such as floppy disks, magnetic tapes, hard disk drives,etc.), optical magnetic storage media (e.g. magneto-optical disks),CD-ROM (Read Only Memory), CD-R, CD-R/W, and semiconductor memories(such as mask ROM, PROM (Programmable ROM), EPROM (Erasable PROM), flashROM, RAM (Random Access Memory), etc.). The program may be provided to acomputer using any type of transitory computer readable media. Examplesof transitory computer readable media include electric signals, opticalsignals, and electromagnetic waves. Transitory computer readable mediacan provide the program to a computer via a wired communication line,such as electric wires and optical fibers, or a wireless communicationline.

First Embodiment

FIG. 1 is a block diagram showing an AD (Analog-to-Digital) converter 1according to a first embodiment. The AD converter 1 shown in FIG. 1 is asemiconductor device having a configuration in which elementsconstituting the circuit are formed on a semiconductor substrate. Thesemiconductor device may include circuits other than the AD converter 1shown in FIG. 1.

The first embodiment illustrates, as an example of the AD converter, anasynchronous-type AD converter that outputs a single conversion resultoutput value (for example, an AD conversion result signal) which isrepresented by a digital value converted from an analog value based onan externally-input input clock signal and performs a conversionoperation based on an internal clock signal which is generated in the ADconverter. The following embodiments can also be applied to asynchronous-type AD converter that is externally supplied with a clocksignal for use in the conversion operation.

As shown in FIG. 1, the AD converter 1 according to the first embodimentincludes a sample-and-hold circuit 10, a digital-to-analog conversioncircuit (for example, a DAC 11), a switch circuit (for example, an inputshort-circuiting switch 12), a comparator 13, and a successiveapproximation control circuit (for example, an asynchronous operationsuccessive approximation logic 14).

The sample-and-hold circuit 10 samples voltage levels of input voltages(for example, a voltage of a positive-side input signal IN+ and avoltage of a negative-side input signal IN−) according to sampling clocksignals (for example, non-overlapping clock signals CK1 and CK2). Thesample-and-hold circuit 10 outputs one of differential voltagescorresponding to the sampled voltage levels to a first comparison line(for example, a positive-side comparison line SND+), and outputs theother of the differential voltages to a second comparison line (forexample, a negative-side comparison line SND−).

The comparator 13 outputs output values (for example, determinationresult signals OUT+ and OUT−) indicating a high level or a low levelaccording to a voltage difference between the first comparison line andthe second comparison line. In the first embodiment, the comparator 13performs an operation for comparing the input differential voltagescorresponding to a determination start signal SRT which is generated bythe asynchronous operation successive approximation logic 14 based on aninput clock signal CKin. Upon completion of the comparison operation,the comparator 13 notifies the asynchronous operation successiveapproximation logic 14 of a determination end signal END.

The comparator 13 receives an offset control value OC. The comparator 13has a function for adjusting an input offset voltage according to theoffset control value OC.

The input short-circuiting switch 12 is provided between the firstcomparison line and the second comparison line. A conductive state and acut-off state of the input short-circuiting switch 12 are switched by aswitch control signal (for example, an input short-circuiting signalSHT) output from the asynchronous operation successive approximationlogic 14.

The DAC 11 changes the voltages of the first comparison line SND+ andthe second comparison line SND− according to the output values (forexample, the determination result signals OUT+ and OUT−). Morespecifically, the DAC 11 according to the first embodiment changes thevoltages of the first comparison line SND+ and the second comparisonline SND− according to a pre-charge signal PC and a DAC control signalCNT which are generated by the asynchronous operation successiveapproximation logic 14 based on the determination result signals OUT+and OUT−.

The asynchronous operation successive approximation logic 14 controlsthe sample-and-hold circuit 11, the comparator 13, and the DAC 11according to a preset successive approximation sequence, and outputs anumber of AD conversion result signals Dout corresponding to the numberof bits of the conversion result output value Dout. Further, uponacquisition of the output value (for example, the determination resultsignal) of all bits of the AD conversion result signals Dout, theasynchronous operation successive approximation logic 14 brings theinput short-circuiting switch 12 into the conductive state, and outputsan offset determination signal (for example, a DCOC comparisondetermination result signal RSLT) for adjusting the input offset voltageof the comparator according to the determination result signal during aperiod in which the input short-circuiting switch 12 is in theconductive state.

The asynchronous operation successive approximation logic 14 generatesthe non-overlapping clock signals CK1 and CK2 and an output clock signalCKout based on the input clock signal CKin. Further, the asynchronousoperation successive approximation logic 14 generates the determinationstart signal SRT indicating a comparison operation timing for thecomparator 13 based on the input clock signal CKin.

An offset control circuit 15 increases or decreases the offset controlvalue OC according to the logic level of the DCOC comparisondetermination result signal RSLT output from the asynchronous operationsuccessive approximation logic 14. This offset control value is a valueindicated by a control signal of a plurality of bits. The offset controlvalue includes the offset control value OC for adjusting the amount ofoffset in a positive direction, and an offset control value OCb foradjusting the amount of offset in a negative direction. In the followingdescription, unless it is necessary to distinguish the offset controlvalue OC from the offset control value OCb, the offset control valueincluding the offset control value OC and the offset control value OCbis denoted by OC.

When the logic level of the DCOC comparison determination result signalRSLT is a first logic level (for example, a high level), the offsetcontrol circuit 15 increases the offset control value OC. When the logiclevel of the DCOC comparison determination result signal RSLT is asecond logic level (for example, a low level), the offset controlcircuit 15 decreases the offset control value OC. In the firstembodiment, the offset control circuit 15 updates the offset controlvalue OC in each conversion cycle in which the AD converter 1 generatesall bit values of the AD conversion result signals Dout. In other words,in the first embodiment, the offset control circuit 15 updates theoffset control value OC every time the AD converter 1 outputs a singleAD conversion result signal Dout.

Next, the sample-and-hold circuit 10, the DAC 11, and the comparator 13will be described in more detail. FIG. 2 is a circuit diagram showingthe sample-and-hold circuit 10 according to the first embodiment.

As shown in FIG. 2, the sample-and-hold circuit 10 includes apositive-side sample-and-hold circuit 21 and a negative-sidesample-and-hold circuit 22. The positive-side sample-and-hold circuit 21samples the positive-side input signal IN+ among ADC input signals, andoutputs the sampled value to the positive-side comparison line SND+ as apositive-side comparison voltage SOUT+. The negative-sidesample-and-hold circuit 22 samples the negative-side input signal IN−among the ADC input signals, and outputs the sampled value to thenegative-side comparison line SND− as a negative-side comparison voltageSOUT−.

The positive-side sample-and-hold circuit 21 includes switches SW1 a,SW1 b, SW2 a, and SW2 b, and sampling capacitors CS1+ and CS2+. Theswitch SW1 a is connected to a node between a first input terminal,which receives the positive-side input signal IN+, and one end of thesampling capacitor CS1+. The switch SW2 a is connected to a node betweenthe first input terminal and one end of the sampling capacitor CS2+. Theswitch SW2 b is connected to a node between one end of the samplingcapacitor CS1+ and a first output terminal connected to thepositive-side comparison line SND+. The switch SW1 b is connected to anode between one end of the sampling capacitor CS2+ and the first outputterminal. The other end of the sampling capacitor CS1+ and the other endof the sampling capacitor CS2+ are connected to each other, and a nodetherebetween is supplied with a reference voltage Vref.

The negative-side sample-and-hold circuit 22 includes switches SW3 a,SW3 b, SW4 a, and SW4 b, and sampling capacitors CS1− and CS2−. Theswitch SW3 a is connected to a node between a second input terminal,which receives the negative-side input signal IN−, and one end of thesampling capacitor CS1−. The switch SW4 a is connected to a node betweenthe second input terminal and one end of the sampling capacitor CS2−.The switch SW4 b is connected to a node between one end of the samplingcapacitor CS1− and a second output terminal connected to thenegative-side comparison line SND−. The switch SW3 b is connected to anode between one end of the sampling capacitor CS2− and the secondoutput terminal. The other end of the sampling capacitor CS1− and theother end of the sampling capacitor CS2− are connected to each other,and a node therebetween is supplied with the reference voltage Vref.

Switching conditions of the switches SW1 a, SW1 b, SW3 a, and SW3 b arecontrolled by the non-overlapping clock signal CK1, and switchingconditions of the switches SW2 a, SW2 b, SW4 a, and SW4 b are controlledby the non-overlapping clock signal CK2. The non-overlapping clocksignals CK1 and CK2 will be described in detail. FIG. 3 shows a timingdiagram for explaining the non-overlapping clock signals used for the ADconverter 1 according to the first embodiment.

As shown in FIG. 3, the non-overlapping clock signals CK1 and CK2 aredifferential clock signals having mutually inverted logic levels. Arising edge of the non-overlapping clock signal CK1 occurs after afalling edge of the non-overlapping clock signal CK2. On the other hand,a falling edge of the non-overlapping clock signal CK1 occurs before arising edge of the non-overlapping clock signal CK2. In other words,when the non-overlapping clock signals CK1 and CK2 are used, a period inwhich two clock signals supplied to the sample-and-hold circuit 10change to the low level is generated. In the example shown in FIG. 3,the two clock signals change to the low level in periods TS2 and TS4.

Next, the operation of the sample-and-hold circuit 10 will be described.FIG. 4 shows a diagram for explaining the operation of thesample-and-hold circuit 10 according to the first embodiment. As shownin FIG. 4, the sample-and-hold circuit 10 repeatedly performs fouroperations. The four operations are carried out in periods TS1 to TS4,respectively, in the timing diagram shown in FIG. 3. FIG. 4 shows thestates of the circuits in each period. Note that in the sample-and-holdcircuit 10, the positive-side sample-and-hold circuit 21 and thenegative-side sample-and-hold circuit 22 operate in the same manner.Accordingly, the operation of only the positive-side sample-and-holdcircuit 21 will now be described.

First, in the period TS1, the non-overlapping clock signal CK1 changesto the high level (for example, a power supply voltage level), and thenon-overlapping clock signal CK2 changes the low level (for example, aground voltage level). Accordingly, in the sample-and-hold circuit 10,the switches SW1 a and SW1 b are in the conductive state and theswitches SW2 a and SW2 b are in the cut-off state. As a result, electriccharge corresponding to the voltage level of the positive-side inputsignal IN+ is accumulated in the sampling capacitor CS1+, and a voltagecorresponding to the amount of electric charge accumulated in thesampling capacitor CS2+ is output as the positive-side comparisonvoltage SOUT+.

In the period TS2, the non-overlapping clock signals CK1 and CK2 changeto the low level. Accordingly, in the sample-and-hold circuit 10, allthe switches SW1 a, SW1 b, SW2 a, and SW2 b are in the cut-off state.This allows the electric charge of each of the sampling capacitors CS1+and CS2+ to be held.

In the period TS3, the non-overlapping clock signal CK1 changes to thelow level and the non-overlapping clock signal CK2 changes to the highlevel. Accordingly, in the sample-and-hold circuit 10, the switches SW1a and SW1 b are in the cut-off state and the switches SW2 a and SW2 bare in the conductive state. As a result, electric charge correspondingto the voltage level of the positive-side input signal IN+ isaccumulated in the sampling capacitor CS2+, and a voltage correspondingto the amount of electric charge accumulated in the sampling capacitorCS1+ is output as the positive-side comparison voltage SOUT+.

In the period TS4, both the non-overlapping clock signals CK1 and CK2change to the low level. Accordingly, in the sample-and-hold circuit 10,all the switches SW1 a, SW1 b, SW2 a, and SW2 b are in the cut-offstate. This allows the electric charge of each of the samplingcapacitors CS1+ and CS2+ to be held.

As seen from the above description, in the sample-and-hold circuit 10, asampling value to be subsequently output to the other sampling capacitoris sampled during a period in which the accumulated sampling value isoutput to one sampling capacitor. This configuration allows thesample-and-hold circuit 10 to reduce the time for switching the samplingperiod and reduce the conversion cycle of the AD converter 1, therebyallowing the AD converter 1 to operate at a higher speed.

Next, the DAC 11 according to the first embodiment will be described indetail. FIG. 5 is a circuit diagram showing the DAC 11 according to thefirst embodiment. As shown in FIG. 5, the DAC 11 according to the firstembodiment has a configuration in which a number of unitdigital-to-analog conversion circuits corresponding to the number ofbits of the AD conversion result signal Dout are connected in parallelbetween DAC output lines DACout+ and DACout−. In the example shown inFIG. 5, the DAC 11 includes unit digital-to-analog conversion circuits301 to 30 n. In this case, the unit digital-to-analog converter 301corresponds to the least significant bit (LSB) of the AD conversionresult signal Dout, and the unit digital-to-analog converter 30 ncorresponds to the most significant bit (MSB) of the AD conversionresult signal Dout.

The unit digital-to-analog converters 301 to 30 n have the sameconfiguration. Accordingly, only the unit digital-to-analog converter 30n will now be described. Note that in FIG. 5, n, n−1, . . . , and 1 (nrepresents the number of bits of the AD conversion result signal Dout)are added to the end of the reference numerals of the respectivecomponents, to thereby represent the components corresponding to therespective bits.

As shown in FIG. 5, the unit digital-to-analog converter 30 n includesswitches 31 n, 32 n, 33 n, 34 n, 35 n, and 36 n, and a pre-chargecapacitor Cdn. One end of the switch 31 n is connected to a power supplyline supplied with a power supply voltage VDD, and the other end of theswitch 31 n is connected to one end of the pre-charge capacitor Cdn. Theother end of the pre-charge capacitor Cdn is connected to one end of theswitch 32 n. The other end of the switch 32 n is connected to a groundline supplied with a ground voltage VSS.

The switch 33 n is connected to a node between the DAC output lineDACout− and a line that connects the switch 31 n and the pre-chargecapacitor Cdn to each other. The switch 34 n is connected to a nodebetween the DAC output line DACout+ and a line that connects the switch32 n and the pre-charge capacitor Cdn to each other. The switch 35 n isconnected to a node between the DAC output line DACout− and the linethat connects the switch 32 n and the pre-charge capacitor Cdn to eachother. The switch 36 n is connected to a node between the DAC outputline DACout+ and the line that connects the switch 31 n and thepre-charge capacitor Cdn to each other. The DAC output line DACout+ isconnected to the positive-side comparison line SND+, and the DAC outputline DACout− is connected to the negative-side comparison line SND−.

Switching conditions of the switches 31 n and 32 n are controlled by thepre-charge signal PC output from the asynchronous operation successiveapproximation logic 14. Switching conditions of the switches 33 n to 36n are controlled by the DAC control signal CNT output from theasynchronous operation successive approximation logic 14. The DACcontrol signal CNT includes a DAC control signal CNTa and a DAC controlsignal CNTb. The DAC control signal CNTa is set to an enabled state (forexample, a high level) when the determination result obtained based onthe determination result signals OUT+ and OUT− is positive. The DACcontrol signal CNTb is set to the enabled state when the determinationresult is negative. The DAC control signals CNTa and CNTb are separatelysupplied to each unit digital-to-analog converter. Thus, in order todistinguish the unit digital-to-analog converters to which the DACcontrol signals are supplied, reference symbols (for example, n, n−1, .. . , and 1) denoting the respective bits of the unit digital-to-analogconverters to which the DAC control signals are supplied are added tothe end of “CNTa” and “CNTb” in FIG. 5. In the example shown in FIG. 5,the switches 33 n and 34 n are each supplied with a DAC control signalCNTan, and the switches 35 n and 36 n are each supplied with a DACcontrol signal CNTbn.

Next, the operation of the DAC 11 according to the first embodiment willbe described. FIG. 6 shows a diagram for explaining the operation of theDAC according to the first embodiment. As shown in FIG. 6, the operationof the DAC 11 can be divided into three periods, i.e., a pre-chargeperiod, a holding period, and an output period. The output period of theDAC 11 can be divided into a positive output and a negative output.

During the pre-charge period, the switches 31 n and 32 n are in theconductive state and the switches 33 n to 36 n are in the cut-off state.This allows the DAC 11 to accumulate electric charge in the pre-chargecapacitor Cdn. Next, during the holding period, the DAC 11 brings theswitches 31 n to 36 n into the cut-off state and holds the electriccharge accumulated in the pre-charge capacitor Cdn.

The DAC 11 provides a positive output or a negative output according tothe result of the determination made in the asynchronous operationsuccessive approximation logic 14. In the positive output, the DAC 11brings the switches 31 n, 32 n, 35 n, and 36 n into the cut-off stateand brings the switches 33 n and 34 n into the conductive state. Thisallows the DAC 11 to decrease the voltage difference between thepositive-side comparison line SND+ and the negative-side comparison lineSND−. On the other hand, in the negative output, the DAC 11 brings theswitches 31 n, 32 n, 33 n, and 34 n into the cut-off state and bringsthe switches 35 n and 36 n into the conductive state. This allows theDAC 11 to increase the voltage difference between the positive-sidecomparison line SND+ and the negative-side comparison line SND−.

Next, the comparator 13 according to the first embodiment will bedescribed in detail. FIG. 7 is a circuit diagram showing the comparator13 according to the first embodiment. As shown in FIG. 7, the comparator13 according to the first embodiment includes a pre-amplifier 40, alatch circuit 41, offset adjustment capacitors CV1 and CV2, and aninverter 42.

The pre-amplifier 40 amplifies the voltage difference between the firstcomparison line (for example, the positive-side comparison line SND+)and the second comparison line (for example, the negative-sidecomparison line SND−), and outputs a first intermediate output voltagepo_m and a second intermediate output voltage po_p. The firstintermediate output voltage po_m is supplied to the latch circuit 41through a first intermediate voltage line. The second intermediateoutput voltage po_p is supplied to the latch circuit 41 through a secondintermediate voltage line.

The offset adjustment capacitor CV1 is connected to the firstintermediate voltage line through which the first intermediate outputvoltage po_m is transmitted. The offset adjustment capacitor CV2 isconnected to the second intermediate voltage line through which thesecond intermediate output voltage po_p is transmitted. The offsetadjustment capacitor CV1 is supplied with the offset control value OCb,and the offset adjustment capacitor CV2 is supplied with the offsetcontrol value OC. The comparator 13 changes the capacitance ratiobetween the offset adjustment capacitor CV1 and the offset adjustmentcapacitor CV2 according to the offset control values OC and OCb, therebyadjusting the input offset voltage. Note that the offset adjustmentcapacitors CV1 and CV2 are capacitors which are formed usingtransistors. The offset adjustment capacitors CV1 and CV2 each have aconfiguration in which a plurality of transistors are connected inparallel, and the capacitance values of the capacitors are changed byswitching between a high-level signal and a low-level signal to besupplied to the source or drain of each transistor according to theoffset control values OC and OCb.

The latch circuit 41 determines the logic level of the output values(for example, the determination result signals OUT+ and OUT−) accordingto the voltage difference between the first intermediate output voltagepo_m and the second intermediate output voltage po_p.

The circuit configurations of the pre-amplifier 40 and the latch circuit41 will now be described in more detail. The pre-amplifier 40 includesPMOS transistors MP0, MP1, and MP2, and NMOS transistors MN1 and MN2.

The gate of the PMOS transistor MP0 is supplied with the determinationstart signal SRT, and the source of the PMOS transistor MP0 is connectedto the power supply line supplied with the power supply voltage VDD. Thedrain of the PMOS transistor MP0 is connected to the sources of the PMOStransistors MP1 and MP2. The PMOS transistor MP0 supplies an operatingcurrent ID to a differential pair composed of the PMOS transistors MP1and MP2 during a period in which the determination start signal SRTsupplied through the inverter 42 is at the low level (at this time, thedetermination start signal SRT is at the high level).

The PMOS transistor MP1 and the NMOS transistor MN1 are connected inseries between the drain of the PMOS transistor MP0 and the ground linesupplied with the ground voltage VSS. The first intermediate outputvoltage po_m is output from a node between the PMOS transistor MP1 andthe NMOS transistor MN1. The gate of the PMOS transistor MP1 isconnected to the positive-side comparison line SND+ and is supplied witha positive input signal (for example, the positive-side comparisonvoltage SOUT+). The gate of the NMOS transistor MN1 is supplied with thedetermination start signal SRT through the inverter 42. The NMOStransistor MN1 is in the cut-off state during a period in which thedetermination start signal SRT supplied through the inverter 42 is atthe low level (at this time, the determination start signal SRT is atthe high level). The NMOS transistor MN1 is in the conductive stateduring a period in which the determination start signal SRT suppliedthrough the inverter 42 is at the high level (at this time, thedetermination start signal SRT is at the low level).

The PMOS transistor MP2 and the NMOS transistor MN2 are connected inseries between the drain of the PMOS transistor MP0 and the ground linesupplied with the ground voltage VSS. The second intermediate outputvoltage po_p is output from a node between the PMOS transistor MP2 andthe NMOS transistor MN2. The gate of the PMOS transistor MP2 isconnected to the negative-side comparison line SND− and is supplied witha negative input signal (for example, the negative-side comparisonvoltage SOUT−). The gate of the NMOS transistor MN2 is supplied with thedetermination start signal SRT through the inverter 42. The NMOStransistor MN2 is in the cut-off state during a period in which thedetermination start signal SRT supplied through the inverter 42 is atthe low level (at this time, the determination start signal SRT is atthe high level). The NMOS transistor MN2 is in the conductive stateduring a period in which the determination start signal SRT suppliedthrough the inverter 42 is at the high level (at this time, thedetermination start signal SRT is at the low level).

The latch circuit 41 includes PMOS transistors MP3 to MP6, NMOStransistors MN3 to MN6, inverters 43 and 44, and a NAND circuit 45.

The sources of the PMOS transistors MP3 to MP6 are each connected to thepower supply line. The gates of the PMOS transistors MP3 and MP4 areeach supplied with the determination start signal SRT. The PMOStransistor MP3 is connected in parallel to the PMOS transistor MP5, andthe PMOS transistor MP4 is connected in parallel to the PMOS transistorMP6.

The PMOS transistor MP5 and the NMOS transistors NM3 and NM5 areconnected in series between the power supply line and the ground line.The NMOS transistor NM3 is provided between the PMOS transistor MP5 andthe NMOS transistor MN5. The gate of the PMOS transistor MP5 and thegate of the NMOS transistor NM5 are connected to each other and are alsoconnected to a line that connects the PMOS transistor MP6 and the NMOStransistor MN4 to each other. The gate of the NMOS transistor MN3receives the first intermediate output voltage po_m.

The PMOS transistor MP6 and the NMOS transistors NM4 and NM6 areconnected in series between the power supply line and the ground line.The NMOS transistor NM4 is provided between the PMOS transistor MP6 andthe NMOS transistor MN6. The gate of the PMOS transistor MP6 and thegate of the NMOS transistor NM6 are connected to each other and are alsoconnected to a line that connects the PMOS transistor MP5 and the NMOStransistor MN3 to each other. The gate of the NMOS transistor MN4receives the second intermediate output voltage po_p.

In the latch circuit 41, a second output signal lo_p is output from theline that connects the PMOS transistor MP5 and the NMOS transistor MN3to each other. The comparator 13 brings the determination result signalOUT− into the low level when the voltage level of the second outputsignal lo_p is higher than a threshold voltage of the inverter 44. Thecomparator 13 brings the determination result signal OUT− into the highlevel when the voltage level of the second output signal lo_p is lowerthan the threshold voltage of the inverter 44.

Further, in the latch circuit 41, a first output signal lo_m is outputfrom the line that connects the PMOS transistor MP6 and the NMOStransistor MN4 to each other. The comparator 13 brings the determinationresult signal OUT+ into the low level when the voltage level of thefirst output signal lo_m is higher than a threshold voltage of theinverter 43. The comparator 13 brings the determination result signalOUT+ into the high level when the voltage level of the first outputsignal lo_m is lower than the threshold voltage of the inverter 43.

Furthermore, in the latch circuit 41, when the voltage differencebetween the first output signal lo_m and the second output signal lo_preaches a sufficient level, the NAND circuit 45 switches the logic levelof the determination end signal END.

Next, the operation of the comparator 13 according to the firstembodiment will be described. In particular, an offset adjustmentoperation and an offset determination process in which the inputshort-circuiting switch 12 causes the positive-side comparison line SND+and the negative-side comparison line SND− to be short-circuited will bedescribed. FIGS. 8 to 10 show timing diagrams for explaining theoperation of the comparator 13 according to the first embodiment. FIG. 8shows a case where the offset control value OC is “0”. FIG. 9 shows acase where the offset control value OC is “+1”. FIG. 10 shows a casewhere the offset control value OC is “+2”. The timing diagrams of FIGS.8 to 10 each show the state in which the input short-circuiting switch12 causes the positive-side comparison line SND+ and the negative-sidecomparison line SND− to be short-circuited, and the voltage differencebetween the signals input to the comparator 13 is zero. In the timingdiagrams of FIGS. 8 to 10, a change in the first intermediate outputvoltage po_m and a change in the second intermediate output voltage po_pduring a period in which the latch circuit 41 determines the magnituderelationship between the first intermediate output voltage po_m and thesecond intermediate output voltage po_p are shown as an enlarged view.

As shown in FIGS. 8 to 10, the comparator 13 causes the firstintermediate output voltage po_m and the second intermediate outputvoltage po_p to rise when the determination start signal SRT changes tothe high level. At this time, in the comparator 13, a difference in therate of rise occurs between the first intermediate output voltage po_mand the second intermediate output voltage po_p due to, for example, avariation in parasitic capacitance depending on the layout, or avariation in the production of semiconductor elements. The comparator 13determines the voltage levels of the first output signal lo_m and thesecond output signal lo_p, which are held in the latch circuit 41, basedon the difference in the rate of rise between the first intermediateoutput voltage po_m and the second intermediate output voltage po_p.Further, the comparator 13 determines the logic levels of thedetermination result signals OUT+ and OUT− according to the voltagelevels of the determined first output signal lo_m and second outputsignal lo_p. Furthermore, upon determination of the logic levels of thedetermination result signals OUT+ and OUT−, the comparator 13 brings thedetermination end signal END into the high level.

During a comparison operation for generating the AD conversion resultsignal Dout having a potential difference between the positive-sidecomparison line SND+ and the negative-side comparison line SND−, adifference in the rate of rise occurs between the first intermediateoutput voltage po_m and the second intermediate output voltage po_pdominantly due to the input voltage difference. However, in a conversionprocess for lower bits in which the potential difference between thepositive-side comparison line SND+ and the negative-side comparison lineSND− decreases, the difference between the offset voltage of thecomparator 13 and the input potential difference decreases, and theoffset voltage has a large effect on the conversion result.

As shown in FIGS. 8 to 10, in the comparator 13, the difference in therate of rise between the first intermediate output voltage po_m and thesecond intermediate output voltage po_p can be reduced by changing theoffset control value OC.

For example, when the case where the offset control value is “0” asshown in FIG. 8 is compared with the case where the offset control valueis “+1” as shown in FIG. 9, the voltage difference between the firstintermediate output voltage po_m and the second intermediate outputvoltage po_p during the determination of the magnitude relationshipbetween the intermediate output voltages of the latch circuit 41 in thetiming diagram shown in FIG. 9 is smaller than that in the timingdiagram shown in FIG. 8.

Further, when the case where the offset control value is “+1” as shownin FIG. 9 is compared with the case where the offset control value is“+2” as shown in FIG. 10, in the timing diagram shown in FIG. 10, themagnitude relationship between the first intermediate output voltagepo_m and the second intermediate output voltage po_p during thedetermination of the magnitude relationship of the intermediate outputvoltages of the latch circuit 41 is reversed and the voltage differencedecreases, as compared with the timing diagram shown in FIG. 9.Accordingly, the determination result in the timing diagram shown inFIG. 10 indicates the logic level inverted from the logic levelindicated by the determination result in the timing diagrams shown inFIGS. 8 and 9.

As seen from the above description, in the comparator 13, the loadcapacitance of the lines from which the first intermediate outputvoltage po_m and the second intermediate output voltage po_p arerespectively output is changed by changing the offset control value OC.This enables the comparator 13 to reduce the input offset voltage bychanging the offset control value OC. As described above, the operationin which two intermediate output voltages are caused to rise insynchronization with the internal clock signal, such as thedetermination start signal SRT, and the magnitude relationship of thevoltage difference between input signals is determined according to thedifference in the rate of rises between the two intermediate outputvoltages is referred to as a dynamic operation.

Next, the operation of the AD converter 1 according to the firstembodiment will be described. FIG. 11 shows a timing diagram forexplaining the operation of the AD converter 1 according to the firstembodiment. As shown in FIG. 11, the AD converter 1 according to thefirst embodiment completes one conversion cycle during one cycle of theexternally-input input clock signal CKin. At this time, the AD converter1 performs an asynchronous operation. Accordingly, the timing at whichthe conversion process is completed during one cycle of the input clocksignal CKin varies depending on circumstances such as the temperature ofthe semiconductor device.

Upon starting the conversion cycle, the AD converter 1 changes the logiclevels of the non-overlapping clock signals CK1 and CK2, and outputs thevalues, which are sampled by the sample-and-hold circuit 10 during theprevious conversion cycle, to the positive-side comparison line SND+ andthe negative-side comparison line SND−. During the period from timing T2to timing T3, the AD converter 1 performs a conversion process fordetermining values from the most significant bit to the leastsignificant bit of the AD conversion result signal Dout by a successiveapproximation process.

In this conversion process, the pulse of the determination start signalSRT is generated at a predetermined cycle, and the comparator 13 outputsthe conversion result signal according to the generated pulse. Upondetermination of the value of the conversion result signal, thecomparator 13 causes the determination end signal END to rise. Theasynchronous operation successive approximation logic 14 determines thelogic level of the DAC control signal CNT in order from the mostsignificant bit side every time the conversion result is obtained. Theasynchronous operation successive approximation logic 14 completes apre-charge operation for the pre-charge capacitors Cdn to Cd1 by thepre-charge signal PC during the period before the start of theconversion process at timing T2.

At timing T3, the asynchronous operation successive approximation logic14 brings the input short-circuiting signal SHT into the high level andbrings the input short-circuiting switch 12 into the conductive state,after all bit values of the AD conversion result signal Dout aredetermined. Further, the asynchronous operation successive approximationlogic 14 outputs a single pulse of the determination start signal SRT inthe state where the positive-side comparison line SND+ and thenegative-side comparison line SND− are short-circuited by the inputshort-circuiting signal SHT, and performs the offset determinationprocess. In the example shown in FIG. 11, a positive determination (forexample, “1”) is made in the offset determination process after timingT3.

At timing T4, the asynchronous operation successive approximation logic14 updates the AD conversion result signal Dout and the DCOC comparisondetermination result signal RSLT, which are obtained in the currentconversion cycle, in response to the start of the subsequent conversioncycle. The offset control circuit 15 updates the offset control value OCaccording to the value of the DCOC comparison determination resultsignal RSLT obtained during the period from timing T1 to timing T4. Thenthe AD converter 1 carries out the conversion process in the conversioncycle after timing T4 in the state where the offset voltage of thecomparator 13 is adjusted based on the offset control value OC updatedat timing T4.

As shown in FIG. 11, in the AD converter 1, a large amount of current isconsumed by the DAC 11, the comparator 13, the asynchronous operationsuccessive approximation logic 14, and the like during the conversionprocess. This causes a phenomenon in which the power supply voltage VDDgradually decreases during the conversion process according to anincrease in current consumption. In this case, the offset voltage of thecomparator 13 has such a feature that the offset voltage changesdepending on a change in the power supply voltage VDD.

Since many of the circuit elements operate during the conversion processin the AD converter 1, the junction temperature of each circuit elementbecomes higher, as the level of bits to be subjected to the conversionprocess becomes lower. In this case, the offset voltage of thecomparator 13 has such a feature that the offset voltage changes due tothe effect of the junction temperature of each circuit element, or theenvironmental temperature of the semiconductor device (thesetemperatures are collectively referred to as a substrate temperature).

In the AD converter 1 according to the first embodiment, the powersupply voltage VDD and the substrate temperature vary during theconversion process as described above. In such a state, the AD converter1 determines the offset voltage immediately after the conversion of theleast significant bit in which the voltage difference to be input to thecomparator 13 is minimized, and then completes the conversion cycle.This allows the AD converter 1 to reduce the offset voltage during theconversion of the least significant bit in which the offset voltage ofthe comparator 13 has a large effect on the conversion result.

The process of updating the offset control value OC in the offsetcontrol circuit 15 according to the first embodiment will now bedescribed. FIG. 12 shows a timing diagram for explaining valuesindicated by the DCOC comparison determination result signal and anoffset adjustment amount in the AD converter.

As shown in FIG. 12, when the DCOC comparison determination resultsignal RSLT obtained in the previous conversion cycle indicates “1” (forexample, positive determination), the offset control circuit 15increases the offset control value OC by 1. On the other hand, when theDCOC comparison determination result signal RSLT obtained in theprevious conversion cycle indicates “0” (for example, negativedetermination), the offset control circuit 15 decreases the offsetcontrol value OC by 1. The offset control value OC is represented byintegers ranging from negative to positive values.

As seen from the above description, in the AD converter 1 according tothe first embodiment, the input short-circuiting switch 12 is providedbetween two lines that supply input signals to the comparator 13.Further, the input short-circuiting switch 12 is brought into theconductive state during the period immediately after the completion ofthe conversion of the least significant bit during one conversion cycle,and the offset voltage of the comparator 13 is determined. In thesubsequent conversion cycle, the conversion process is carried out bythe comparator 13 whose offset voltage has been corrected based on theresult of the determination of the offset voltage.

This operation allows the AD converter 1 according to the firstembodiment to reduce the offset voltage during the conversion of theleast significant bit in which the offset voltage of the comparator 13has a large effect on the conversion result. Moreover, the reduction inthe offset voltage during the conversion of the least significant bitenables the AD converter 1 according to the first embodiment to achievea high conversion accuracy.

While the comparator 13 that performs the dynamic operation has beendescribed above, the offset correction method can also be applied to aconverter that performs a static operation. For example, if inputsignals are set to the same potential by a switch provided at the inputterminal of the converter immediately after the conversion of the leastsignificant bit as described above, the converter that performs thestatic operation can also obtain a conversion result with high accuracy.

The asynchronous-type AD converter that performs the conversionoperation for each bit independently of the cycle of theexternally-input input clock signal CKin has been described above. Inthis asynchronous-type AD converter, as described above, the frequencyof the clock signal, which is generated by the asynchronous operationsuccessive approximation logic 14 and indicates the conversion timingfor each bit, varies depending on the substrate temperature. This makesit difficult to determine the conversion timing of the least significantbit. However, in the AD converter 1 according to the first embodiment,the asynchronous operation successive approximation logic 14 thatcontrols the conversion timing for each bit outputs the inputshort-circuiting signal SHT for controlling the switching condition ofthe input short-circuiting switch 12. Accordingly, even when theconversion timing of the least significant bit is indefinite as in theasynchronous-type AD converter, the offset voltage of the comparator 13can be determined immediately after the conversion of the leastsignificant bit.

As described above, in the AD converter 1 according to the firstembodiment, an extremely small offset voltage that affects theconversion result of the least significant bit is corrected.Accordingly, in order to perform the offset correction processaccurately, it is necessary to prevent a variation in the parasiticcapacitance or the like, which is parasitic to the comparator 13 and theline through which the input voltage of the comparator 13 istransmitted, from occurring between the positive-side comparison lineSND+ and the negative-side comparison line SND−. In this regard, thelayout associated with the comparator 13 of the AD converter 1 accordingto the first embodiment will be described below.

First, FIG. 13 shows a schematic diagram for explaining the layout ofthe comparator 13 according to the first embodiment. As shown in FIG.13, the comparator 13 has a configuration in which a comparator bodyportion and the offset adjustment capacitors CV1 and CV2 are arranged ina divided manner. The comparator body portion includes the pre-amplifier40, the latch circuit 41, and the inverter 42 which are shown in FIG. 7.The input short-circuiting switch 12 is formed in a region which issandwiched between the comparator body portion and the offset adjustmentcapacitors CV1 and CV2.

As shown in FIG. 13, the positive-side comparison line SND+ and thenegative-side comparison line SND− are formed in an upper layer of theinput short-circuiting switch 12 and are arranged between the comparatorbody portion and the offset adjustment capacitors CV1 and CV2. Thepositive-side comparison line SND+ and the negative-side comparison lineSND− are formed in a region extending from one end of the region inwhich the comparator 13 is formed to the other end thereof. In theexample shown in FIG. 13, the length of a side at which the offsetadjustment capacitors CV1 and CV2 are parallel to the positive-sidecomparison line SND+ and the negative-side comparison line SND− islonger than the length of a side at which the comparator body isparallel to the positive-side comparison line SND+ and the negative-sidecomparison line SND−. Accordingly, in the example shown in FIG. 13, thepositive-side comparison line SND+ and the negative-side comparison lineSND− are formed so as to have at least the length from one end of theregion in which the offset adjustment capacitors CV1 and CV2 are formedto the other end thereof. Note that in FIG. 13, only the main lineportions of the positive-side comparison line SND+ and the negative-sidecomparison line SND− are shown. At one end of each of the positive-sidecomparison line SND+ and the negative-side comparison line SND−, thesample-and-hold circuit 10 or the DAC 11 is disposed.

The arrangement of the positive-side comparison line SND+ and thenegative-side comparison line SND− in such a manner that the lines arenot stopped at the node between the input short-circuiting switch 12 andthe lines makes it possible to form the two lines with the same lengthand to reduce the difference in parasitic capacitance between the lines.Further, the comparator body portion and the offset adjustmentcapacitors CV1 and CV2 are arranged in a divided manner, and thepositive-side comparison line SND+, the negative-side comparison lineSND−, and the input short-circuiting switch 12 are arranged between thecomparator body portion and the offset adjustment capacitors. Thisarrangement makes it possible to reduce the distance at which thepositive-side comparison line SND+ and the negative-side comparison lineSND− are parallel to the region in which the circuit elementsconstituting the comparator 13 are formed, and to reduce the parasiticcapacitance.

As shown in FIG. 13, in the comparator 13, shield lines 52 are formed soas to sandwich the positive-side comparison line SND+ and thenegative-side comparison line SND−. The arrangement of the shield lines52 in this manner leads to a reduction in the effect of noise, which iscaused due to operations of other circuit elements, on the voltagelevels of the positive-side comparison line SND+ and the negative-sidecomparison line SND−.

As shown in FIG. 13, in the comparator 13, an input short-circuitingswitch control line 51 is formed in a region other than the region inwhich the comparator body portion and the offset adjustment capacitorsCV1 and CV2 are formed. In the example shown in FIG. 13, the inputshort-circuiting switch control line 51 is formed in a region betweenthe comparator body portion and the offset adjustment capacitors CV1 andCV2. Further, the input short-circuiting switch control line 51 isformed so as to have the length from one end of the region in which thecomparator 13 is formed to the other end thereof, as with thenegative-side comparison line SND− and the positive-side comparison lineSND+. This arrangement allows the parasitic capacitance, which is formeddue to the input short-circuiting switch control line 51, to beuniformly formed in the offset adjustment capacitor CV1 and the offsetadjustment capacitor CV2, and leads to a reduction in the differencebetween the capacitances of the offset adjustment capacitors CV1 andCV2. Furthermore, this arrangement makes it possible to reduce theeffect of the parasitic capacitance of the input short-circuiting switchcontrol line 51 on the circuit elements in the comparator body portion.

The input short-circuiting switch control line 51 is formed in parallelto the negative-side comparison line SND− with the shield line 52interposed therebetween. This arrangement makes it possible to reducethe parasitic capacitance formed due to the input short-circuitingswitch control line 51, and prevents unevenness in the parasiticcapacitance.

As shown in FIG. 13, the comparator 13 further includes capacitorconnecting lines 53 which are formed so as to connect the comparatorbody portion and the offset adjustment capacitors with the inputshort-circuiting switch 12, the positive-side comparison line SND+, thenegative-side comparison line SND−, the input short-circuiting switchcontrol line 51, and the shield lines 52 interposed therebetween, andwhich connect comparator body internal lines, which are formed withinthe comparator body, and the offset adjustment capacitors to each other.The capacitor connecting lines 53 are arranged so as to be orthogonal tothe positive-side comparison line SND+, the negative-side comparisonline SND−, the input short-circuiting switch control line 51, and theshield lines 52. This arrangement makes it possible to reduce theparasitic capacitance between the capacitor connecting lines 53 andother lines, such as the positive-side comparison line SND+, and tosuppress a variation in the parasitic capacitance between other lines.

As shown in FIG. 13, in the comparator 13, the positive-side comparisonline SND+, the negative-side comparison line SND−, and the shield lines52 are formed in the same wiring layer (for example, a fifth wiringlayer M5). Further, in the comparator 13, the positive-side andnegative-side comparison lines SND+ and SND− and the inputshort-circuiting switch control line 51 are formed in different wiringlayers which may be formed with at least one wiring layer or no wiringlayer interposed therebetween. In the example shown in FIG. 13, theinput short-circuiting switch control line 51 is formed in a firstwiring layer Ml. Furthermore, in the comparator 13, the positive-sideand negative-side comparison lines SND+ and SND− and the capacitorconnecting lines 53 are formed in different wiring layers which may beformed with at least one wiring layer or no wiring layer interposedtherebetween. As for the capacitor connecting lines 53, it is onlynecessary that at least portions at which the capacitor connecting lines53 are orthogonal to the positive-side comparison line SND+ and thenegative-side comparison line SND− are formed in different wiringlayers. In the example shown in FIG. 13, the capacitor connecting lines53 are formed in a third wiring layer M3. This arrangement makes itpossible to further suppress the parasitic capacitance between lines.

Next, the layout of the input short-circuiting switch 12, thepositive-side comparison line SND+, the negative-side comparison lineSND−, and the comparator 13 will be described in more detail. FIG. 14shows a schematic diagram for explaining the layout of the comparator,the switch circuit, and the lines according to the first embodiment. InFIG. 14, only the PMOS transistors MP1 and MP2 which are supplied withinput signals in the comparator body portion are illustrated as thecomparator body portion.

In the example shown in FIG. 14, the PMOS transistor MP1 corresponds toa first transistor having a gate connected to the first comparison line(for example, the positive-side comparison line SND+), and the PMOStransistor MP2 corresponds to a second transistor having a gateconnected to the second comparison line (for example, the negative-sidecomparison line SND−).

As shown in FIG. 14, the input short-circuiting switch 12 is formed soas to be adjacent to both a first transistor region in which the PMOStransistor MP1 is formed and a second transistor region in which thePMOS transistor MP2 is formed.

In the comparator 13, a distance W1 at which a switch transistor regionin which a switch transistor constituting the input short-circuitingswitch 12 is formed is parallel to the first transistor region is set tobe substantially equal to a distance W2 at which the second transistorregion and the switch transistor region are parallel to each other.Further, in the comparator 13, the distance W1 at which the switchtransistor region is parallel to a region in which the offset adjustmentcapacitor CV1 is formed is set to be substantially equal to the distanceW2 at which the switch transistor region is parallel to a region inwhich the offset adjustment capacitor CV2 is formed. This arrangementmakes it possible to suppress the effect of the input short-circuitingswitch 12 on variations in right and left circuit configurations of thecomparator body portion and the offset adjustment capacitors CV1 andCV2.

Furthermore, as shown in FIG. 14, the positive-side comparison line SND+includes a main line 54 and a branch line 56, and the negative-sidecomparison line SND− includes a main line 55 and a branch line 56. Themain lines 54 and 55 pass through the upper layer of the switch circuitand are formed in a region extending from one end of the region in whichthe comparator is formed to the other end thereof. The branch line 56branches from the main line 54 and connects the main line with the gateof the PMOS transistor MP1. The branch line 57 branches from the mainline 55 and connects the main line with the gate of the PMOS transistorMP2. Each main line and each branch line are connected via a via-hole.The branch lines 56 and 57 are formed in a wiring layer different fromthat of the main lines, and are formed with substantially the samelength in the direction in which the main lines and the gates of thetransistors are respectively connected to each other. In the exampleshown in FIG. 14, a length L1 of the branch line 56 and a length L2 ofthe branch line 57 are set to be equal to each other.

If the branch lines are formed simply based on the distance betweennodes, a difference in length occurs between the two branch lines. Thisdifference in length causes a variation in the parasitic capacitance.However, as shown in FIG. 14, the lengths of the branch lines are set tobe equal to each other, regardless of the distance between nodes,thereby suppressing the effect of the length of each branch line on avariation in the parasitic capacitance.

Next, the arrangement relationship the positive-side and negative-sidecomparison lines SND+ and SND− and the input short-circuiting switch 12will be described. FIG. 15 shows a schematic diagram for explaining indetail the layout of the switch circuit and the comparison linesaccording to the first embodiment.

As shown in FIG. 15, in the AD converter 1 according to the firstembodiment, the positive-side comparison line SND+ and the negative-sidecomparison line SND− are arranged so as to sandwich a center point CDFof a diffusion region which serves as the source or drain of thetransistor that constitutes the input short-circuiting switch 12.Assuming that the width of the diffusion region is represented by W3 andthe length of the diffusion region is represented by W4, for example,the center point CDF can be defined as an intersection of centerlines ofthe width W3 and the width W4, or as an intersection of diagonal linesof the diffusion region. This arrangement prevents a variation in theparasitic capacitance, which is formed between the inputshort-circuiting switch 12 and the positive-side and negative-sidecomparison lines SND+ and SND−, between the two lines.

In another aspect, in the AD converter 1 according to the firstembodiment, the positive-side comparison line SND+ and the negative-sidecomparison line SND− are arranged so as to sandwich a center point CG inthe gate width direction of the gate of the transistor constituting theinput short-circuiting switch 12. This arrangement prevents a variationin the parasitic capacitance, which is formed between the inputshort-circuiting switch 12 and the positive-side and negative-sidecomparison lines SND+ and SND−, between the two lines.

As shown in FIG. 15, in the AD converter 1 according to the firstembodiment, contacts CTa that connect the positive-side comparison lineSND+ and the transistor of the input short-circuiting switch 12 to eachother, and contacts CTb that connect the negative-side comparison lineSND− and the transistor of the input short-circuiting switch 12 to eachother are provided so as to sandwich the gate of the transistor of theinput short-circuiting switch 12. The number of contacts CTa is set tobe equal to the number of contacts CTb. In the input short-circuitingswitch 12, the number of diffusion regions to which the positive-sidecomparison line SND+ is connected is set to be equal to the number ofdiffusion regions to which the negative-side comparison line SND− isconnected. This arrangement prevents a variation in the parasiticcapacitance, which is formed between the input short-circuiting switch12 and the positive-side and negative-side comparison lines SND+ andSND−, between the two lines.

As described above, in the AD converter 1 according to the firstembodiment, a variation in the parasitic capacitance between two pathsfor the comparison operation is suppressed by devising the layout of thecircuit elements of the comparator 13, the input short-circuiting switch12, the positive-side comparison line SND+, and the negative-sidecomparison line SND−. Thus, a variation in the parasitic capacitancebetween two paths is suppressed, thereby making it possible to determinea small offset voltage without error. In particular, in the AD converter1 according to the first embodiment, the offset voltage is determinedimmediately after the comparison operation for the least significant bitis carried out to compare input voltages having an extremely smallvoltage difference. Therefore, it is extremely important to suppress avariation between paths in this manner so as to suppress an erroneousdetermination.

Second Embodiment

The first embodiment illustrates an example in which the offset controlvalue OC is updated in each conversion cycle. On the other hand, asecond embodiment illustrates an example in which offset determinationresults (values indicated by the DCOC comparison determination resultsignal RSLT) in a plurality of conversion cycles are integrated and theoffset control value is updated based on a ratio between the number ofpositive determinations included in the integrated value and the numberof negative determinations included in the integrated value.

FIG. 16 shows a timing diagram for explaining the values indicated bythe DCOC comparison determination result signal and the offsetadjustment amount in the AD converter 1 according to the secondembodiment. As shown in FIG. 16, in the second embodiment, the offsetcontrol circuit 15 updates the offset control value when the number ofconversion cycles in which the output values of all bits of theconversion result output value (for example, the AD conversion resultsignal Dout) are generated reaches a preset number of cycles.

FIG. 17 is a graph for explaining the values indicated by the DCOCcomparison determination result signal and the offset adjustment amountin the offset control circuit 15 according to the second embodiment. Inthe example shown in FIG. 17, the offset control value OC is increasedwhen the number of positive determinations is greater than 50% of theratio between the number of positive determinations and the number ofnegative determinations included in the integrated value of the DCOCcomparison determination result signal RSLT, and the offset controlvalue OC is decreased when the number of negative determinations isgreater than 50% of the ratio.

FIG. 18 shows another example showing the criteria for updating theoffset control value OC. In the example shown in FIG. 18, the offsetcontrol value OC is increased when the ratio of the number of positivedeterminations included in the integrated value of the DCOC comparisondetermination result signal RSLT is equal to or greater than 75%; theoffset control value OC is decreased when the ratio of the number ofpositive determinations is equal to or less than 25%; and the offsetcontrol value OC is maintained when the ratio between the number ofpositive determinations and the number of negative determinations is inthe range from 25% to 75%.

In this manner, the values indicated by the DCOC comparisondetermination result signal RSLT obtained in a plurality of conversioncycles are integrated and the offset control value OC is updated basedon the ratio between the number of positive determinations and thenumber of negative determinations included in the integrated value,thereby achieving a more stable operation than when the offset controlvalue OC is updated in each conversion cycle.

In this regard, various forms of the number of conversion cycles to beintegrated can be used. FIGS. 19 to 21 are graphs for explaining therelationship between the update timing of the offset control value andthe number of integrations of the values indicated by the DCOCcomparison determination result signal in the AD converter according tothe second embodiment.

In the example shown in FIG. 19, the number of processing cycles ofintegration is set to be constant, regardless of an update timing “i” ofthe offset control value OC. In the example shown in FIG. 20, the numberof integrations is changed according to the update timing “i” of theoffset control value OC. In the example shown in FIG. 20, the number ofintegrations is increased or decreased in a constant cycle and within aconstant range. In the example shown in FIG. 21, the number ofintegrations is randomly changed for each update timing “i” of theoffset control value OC. In the case of randomly changing the number ofintegrations, a pseudorandom generator that generates random numbers byusing a middle-square method, a linear congruential method, a linearfeedback shift register, or the like can be used.

In this manner, when the number of conversion cycles to be integrated isset to be constant as shown in FIG. 19, noise (for example, spurioustones) depending on the update cycle may be generated. However, as shownin FIGS. 20 and 21, for example, when the number of conversion cycles tobe integrated is changed, a fluctuation occurs in the integrated value,which makes it possible to suppress spurious tones.

The invention made by the present inventor has been described in detailabove with reference to embodiments. However, the present invention isnot limited to the embodiments described above, and can be modified invarious manners without departing from the scope of the invention.Further, the scope of the claims is not limited by the embodimentsdescribed above. Furthermore, it is noted that, Applicant's intent is toencompass equivalents of all claim elements, even if amended laterduring prosecution.

For example, in the semiconductor device according to the embodimentsdescribed above, the conductivity type (p-type or n-type) of thesemiconductor substrate, semiconductor layer, diffusion layer (diffusionregion), and the like may be reversed. Accordingly, when one of theconductivity types of the n-type and the p-type is defined as a firstconductivity type and the other conductivity type is defined as a secondconductivity type, the first conductivity type may be the p-type and thesecond conductivity type may be the n-type. On the contrary, the firstconductivity type may be the n-type and the second conductivity type maybe the p-type.

The first and second embodiments can be combined as desirable by one ofordinary skill in the art.

What is claimed is:
 1. A semiconductor device comprising: asample-and-hold circuit that samples a voltage level of an input voltageaccording to a sampling clock signal, outputs one of differentialvoltages corresponding to the voltage level of the input voltage to afirst comparison line, and outputs the other of the differentialvoltages to a second comparison line; a comparator that outputs anoutput value indicating a high level or a low level according to avoltage difference between the first comparison line and the secondcomparison line; a digital-to-analog conversion circuit that changesvoltages of the first comparison line and the second comparison lineaccording to the output value; a successive approximation controlcircuit that controls the sample-and-hold circuit, the comparator, andthe digital-to-analog conversion circuit according to a presetsuccessive approximation sequence, and outputs a number of output valuescorresponding to the number of bits of a conversion result output value;and a switch circuit that is provided between the first comparison lineand the second comparison line, a conductive state and a cut-off stateof the switch circuit being switched by the successive approximationcontrol circuit, wherein upon acquisition of the output value of allbits of the conversion result output value, the successive approximationcontrol circuit brings the switch circuit into the conductive state, andoutputs an offset determination signal for adjusting an input offsetvoltage of the comparator according to the output value during a periodin which the switch circuit is in the conductive state, and thecomparator changes the input offset voltage according to an offsetcontrol value generated according to a logic level of the offsetdetermination signal.
 2. The semiconductor device according to claim 1,further comprising an offset control circuit that increases or decreasesthe offset control value according to the logic level of the offsetdetermination signal output from the successive approximation controlcircuit, wherein the comparator includes: a pre-amplifier that amplifiesthe voltage difference between the first comparison line and the secondcomparison line, and outputs a first intermediate output voltage and asecond intermediate output voltage; a latch circuit that determines alogic level of the output value according to a voltage differencebetween the first intermediate output voltage and the secondintermediate output voltage; a first offset adjustment capacitorconnected to a first intermediate voltage line through which the firstintermediate output voltage is transmitted; and a second offsetadjustment capacitor connected to a second intermediate voltage linethrough which the second intermediate output voltage is transmitted, acapacitance ratio between the first offset adjustment capacitor and thesecond offset adjustment capacitor being changed according to the offsetcontrol value.
 3. The semiconductor device according to claim 2, whereinwhen the logic level of the offset determination signal is a first logiclevel, the offset control circuit increases the offset control value,and when the logic level of the offset determination signal is a secondlogic level, the offset control circuit decreases the offset controlvalue.
 4. The semiconductor device according to claim 2, wherein theoffset control circuit updates the offset control value in eachconversion cycle in which all bit values of the conversion result outputvalue are generated.
 5. The semiconductor device according to claim 2,wherein the offset control circuit updates the offset control value whenthe number of conversion cycles in which output values of all bits ofthe conversion result output value are generated reaches a preset numberof cycles.
 6. The semiconductor device according to claim 5, wherein theoffset control circuit changes the number of cycles every time theoffset control circuit updates the offset control value.
 7. Thesemiconductor device according to claim 5, wherein the offset controlcircuit determines a direction in which the offset control value ischanged, according to a ratio between the number of positivedeterminations and the number of negative determinations, the positivedeterminations and the negative determinations being included in anintegrated value of values indicated by the offset determination signalcorresponding to the preset number of cycles.